Method and apparatus for decomposing channel in closed-loop multiple input multiple output communication system

ABSTRACT

An apparatus for decomposing a channel in a closed-loop Multiple Input Multiple Output (MIMO) communication system is provided. The channel decomposition apparatus includes a transmitter for preceding input symbols using a first matrix which is a product of a unitary matrix V, a diagonal matrix Φ and a blockwise Uniform Channel Decomposition (UCD) matrix P BL , before outputting the input symbols.

CROSS-REFERENCE TO RELATED APPLICATION(S) AND CLAIM OF PRIORITY

The present application claims the benefit under 35 U.S.C. §119(a) of a Korean Patent Application filed in the Korean Intellectual Property Office on Nov. 9, 2007 and assigned Serial No. 2007-114286, the disclosures of which are incorporated herein by reference.

TECHNICAL FIELD OF THE INVENTION

The present invention relates to a method and apparatus for decomposing all channels into independent subchannels in a closed-loop Multiple Input Multiple Output (MIMO) communication system.

BACKGROUND OF THE INVENTION

In communication systems, various schemes for acquiring high performance gain are being studied, and their typical examples include an open-loop scheme, a closed-loop scheme, a multi-antenna scheme, and a closed-loop multi-antenna scheme.

For example, compared with the open-loop system in which only the receiver can use channel information, the closed-loop system in which both the transmitter and the receiver can use channel information can have a higher performance gain.

In addition, the use of multiple antennas can also provide greater performance gain. The multi-antenna system is roughly classified into a method for obtaining diversity gain to decrease a transmission error rate and a multiplexing method for simultaneously transmitting many symbols to increase a data rate.

Based on these methods, in order to secure higher system performance, research is being conducted on a closed-loop multi-antenna technology capable of obtaining both diversity gain and multiplexing gain by using multiple antennas for the closed-loop system in which the transmitter can acquire channel information.

The typical closed-loop multi-antenna technology includes a Singular Value Decomposition (SVD) technique and a Uniform Channel Decomposition (UCD) technique. The SVD technique decomposes all channels for multiple antennas into independent subchannels, so the independent subchannels have various Signal-to-Noise Ratios (SNRs). However, in the SVD technique, low-SNR subchannels suffer performance reduction, causing an increase in the performance imbalance of the entire system.

In order to solve the problems of the SVD technique, the UCD technique was developed. The UCD technique achieves the same subchannel SNR for each subcarrier by applying a Successive Interference Cancellation (SIC) method to the receiver. Thus, compared with the SVD technique, the UCD technique noticeably improves error performance when there is no channel coding.

On the other hand, during channel coding, the UCD technique may undergo abrupt performance reduction due to an error transfer phenomenon that inevitably occurs with the SIC method used by the receiver.

SUMMARY OF THE INVENTION

To address the above-discussed deficiencies of the prior art, it is a primary aspect of the present invention to address at least the problems and/or disadvantages and to provide at least the advantages described below. Accordingly, an aspect of the present invention is to provide a blockwise transmission precoder for increasing the performance reduced due to an error transfer phenomenon of UCD by using a structure of the optimal Maximum Likelihood (ML) receiver which is free from the error transfer phenomenon.

Another aspect of the present invention is to provide a method and apparatus for enabling simple symbol-by-symbol detection at a receiver by means of a transmission precoder, and also acquiring optimal performance of a multi-antenna system.

According to one aspect of the present invention, there is provided an apparatus for decomposing a channel in a closed-loop Multiple Input Multiple Output (MIMO) communication system. The apparatus includes a transmitter for preceding input symbols using a first matrix which is a product of a unitary matrix V, a diagonal matrix Φ and a blockwise Uniform Channel Decomposition (UCD) matrix P_(BL), before outputting the input symbols.

According to one aspect of the present invention, there is provided a method for decomposing a channel in a closed-loop Multiple Input Multiple Output (MIMO) communication system. The method includes precoding input symbols using a first matrix which is a product of a unitary matrix V, a diagonal matrix Φ and a blockwise Uniform Channel Decomposition (UCD) matrix P_(BL), before outputting the input symbols.

Before undertaking the DETAILED DESCRIPTION OF THE INVENTION below, it may be advantageous to set forth definitions of certain words and phrases used throughout this patent document: the terms “include” and “comprise,” as well as derivatives thereof, mean inclusion without limitation; the term “or,” is inclusive, meaning and/or; the phrases “associated with” and “associated therewith,” as well as derivatives thereof, may mean to include, be included within, interconnect with, contain, be contained within, connect to or with, couple to or with, be communicable with, cooperate with, interleave, juxtapose, be proximate to, be bound to or with, have, have a property of, or the like. Definitions for certain words and phrases are provided throughout this patent document, those of ordinary skill in the art should understand that in many, if not most instances, such definitions apply to prior, as well as future uses of such defined words and phrases.

BRIEF DESCRIPTION OF THE DRAWINGS

For a more complete understanding of the present disclosure and its advantages, reference is now made to the following description taken in conjunction with the accompanying drawings, in which like reference numerals represent like parts:

FIGS. 1A and 1B are diagrams illustrating structures of a transmitter and a receiver according to an embodiment of the present invention;

FIG. 2 is a diagram illustrating a structure of a soft demapper according to an embodiment of the present invention;

FIG. 3 is a diagram illustrating an example of an error rate versus SNR of BL-UCD according to an embodiment of the present invention; and

FIG. 4 is a diagram illustrating another example of an error rate versus SNR of BL-UCD according to an embodiment of the present invention.

DETAILED DESCRIPTION OF THE INVENTION

FIGS. 1A through 4, discussed below, and the various embodiments used to describe the principles of the present disclosure in this patent document are by way of illustration only and should not be construed in any way to limit the scope of the disclosure. Those skilled in the art will understand that the principles of the present disclosure may be implemented in any suitably arranged communication system.

FIGS. 1A and 1B are diagrams illustrating structures of a transmitter and a receiver according to an embodiment of the present invention. Herein, a closed-loop system between a transmitter 100 having N_(t) transmit antennas and a receiver 120 having N_(r) receive antennas is considered, and the transmitter 100 and the receiver 120 are assumed to acquire perfect channel state information of each other.

Referring to FIG. 1A, the transmitter 100 includes a convolution encoder 102, a bit interleaver 104, a serial-parallel converter 106, mapper modems 108-1˜108-N, a precoder 110, and transmit antennas 112-1˜112-N.

Input bits are subjected to Bit Interleaved Coded Modulation (BICM)-based channel encoding through the convolution encoder 102 and the bit interleaver 104, and then input to the mapper modems 108-1˜108-N where an N^(th)-order complex symbol vector S shown as Equation 1 below is generated and then input to the precoder 110.

The convolution encoder 102 supports all of N transmit antennas 112-1˜112-N. As a result, Blockwise-Uniform Channel Decomposition (BL-UCD) is applied by making a subblock by grouping 2 subchannels of the highest-SNR subchannel and the lowest-SNR subchannel by means of the precoder 110 and a receive filter unit. A detailed description of BL-UCD will be given below.

s=[s₁s₂ . . . s_(N)]^(T).   [Eqn. 1]

The precoder 110 precodes the N^(th)-order complex symbol vector s, or an input signal, using an N×N_(t) precoder matrix F, thereby outputting an N_(t) ^(th)-order complex transmit signal vector x shown as Equation 2:

x=[x₁x₂ . . . x_(n) ₁ ]^(T).   [Eqn. 2]

Referring to FIG. 1B, the receiver 120 includes N_(r) receive antennas 122-1˜122-N, a receive filter unit 124, a soft demapper 126, parallel-serial converter 128, a bit deinterleaver 130, and a Viterbi decoder 132.

The N_(r) receive antennas 122-1˜122-N receive an N_(r) ^(th)-order complex reception signal vector y shown as Equation 3 below, and transmit it to the receive filter unit 124.

The receive filter unit 124 outputs a signal vector {tilde over (y)} obtained by performing reception filtering on the y using a receive filter G to which BL-UCD is applied. For each block, the soft demapper 126 independently acquires real parts and imaginary parts of elements constituting the s using the F and the G for the {tilde over (y)}. Since the remaining structures are not closely related to the present invention, a detailed description thereof will be omitted.

$\begin{matrix} {\begin{matrix} {y = \left\lbrack {y_{1}y_{2}\mspace{14mu} \ldots \mspace{14mu} y_{N_{r}}} \right\rbrack^{T}} \\ {{= {{{Hx} + w} = {{HFs} + w}}},} \end{matrix}\begin{matrix} {y = \left\lbrack {y_{1}y_{2}\mspace{14mu} \ldots \mspace{14mu} y_{N_{r}}} \right\rbrack^{T}} \\ {{= {{{Hx} + w} = {{HFs} + w}}},} \end{matrix}} & \left\lbrack {{Eqn}.\mspace{14mu} 3} \right\rbrack \end{matrix}$

where H denotes an N_(t)×N_(r) channel matrix.

According to the present invention, a detailed description will now be made of a BL-UCD method to which blockwise ML detection is applied to prevent the error transfer phenomenon.

In order to increase the minimum gain value of each subchannel, the precoder 110 makes subblocks by grouping 2 subchannels of the highest-SNR subchannel and the lowest-SNR subchannel for all subchannels, and then precodes them using the UCD technique.

Specifically, the precoder 110 matches the number N of non-zero singular values to the number K of transmitted streams, and reorders the descending-ordered singular values of a channel matrix H based on the SVD technique, as shown in Equation 4:

$\begin{matrix} {{H = {{U\; \Lambda \; V^{\dagger}}\overset{\Delta}{=}{\left\lbrack {u_{1}u_{2}\mspace{14mu} \ldots \mspace{14mu} u_{K}} \right\rbrack {\Lambda \left\lbrack {v_{1}v_{2}\mspace{14mu} \ldots \mspace{14mu} v_{K}} \right\rbrack}^{\dagger}}}},} & \left\lbrack {{Eqn}.\mspace{14mu} 4} \right\rbrack \end{matrix}$

where u and v are each a unitary matrix, and a diagonal matrix Λ having a singular value λ_(i) as a diagonal element is Λ=diag{λ₁,λ_(N),λ₂,λ_(N-1), . . . ,λ_(N/2),λ_(N/2+1)}. The singular valves λ_(i) are numbered according to their sizes (λ₁≧λ₂≧ . . . ≧λ_(N)).

Thereafter, the precoder 110 in the transmitter 100 precodes output signals of the mapper modems 108-1˜108-N using a precoder matrix F defined as Equation 5:

F=VΦP_(BL),   [Eqn. 5]

where Φ denotes an N×N diagonal matrix having power loading parameters and is diag{φ₁,φ_(N),φ₂,φ_(N-1), . . . ,φ_(N/2),φ_(N/2+1)}, and P_(BL) denotes a unitary matrix.

The Φ can be a unit matrix when power loading is not applied, and can be found by the well-known water-filling solution using Equation 6:

$\begin{matrix} {{\varphi_{k} = \left( {\mu - \frac{\alpha}{\lambda_{k}^{2}}} \right)_{+}^{\frac{1}{2}}},} & \left\lbrack {{Eqn}.\mspace{14mu} 6} \right\rbrack \end{matrix}$

where μ is set to satisfy Σ_(k=1) ^(N)φ_(k) ²=N, α denotes an energy ratio of a transmission symbol to noise, and

$(a)_{+}\overset{\Delta}{=}{\max {\left\{ {0,a} \right\}.}}$

Here, N is found through a BL-UCD process based on Equation 7:

$\begin{matrix} {{P_{BL}\overset{\Delta}{=}{{diag}\left\{ {P_{1},P_{2},\ldots \mspace{14mu},P_{\frac{N}{2}}} \right\}}},} & \left\lbrack {{Eqn}.\mspace{14mu} 7} \right\rbrack \end{matrix}$

where P_(i) denotes a preceding matrix of an i^(th) subblock, and

${P_{i} = {{\begin{bmatrix} C_{i} & S_{i} \\ {- S_{i}} & C_{i} \end{bmatrix}\mspace{14mu} {for}\mspace{14mu} i} = 1}},2,\ldots \mspace{14mu},{N/2.}$

Here,

$C_{i} = {\sqrt{\frac{\sqrt{\left( {\sigma_{i}^{2} + \alpha} \right)\left( {\sigma_{N - i + 1}^{2} + \alpha} \right)} - \left( {\sigma_{N - i + 1}^{2} + \alpha} \right)}{\sigma_{i}^{2} - \sigma_{N - i + 1}^{2}}}\mspace{14mu} {and}}$ ${S_{i} = \sqrt{1 - C_{i}^{2}}},{{{where}\mspace{14mu} \sigma_{i}} = {\lambda_{i}{\varphi_{i}.}}}$

Meanwhile, the receive filter unit 124 in the receiver 120 outputs a signal {tilde over (y)} obtained by performing reception filtering on an N_(r) ^(th)-order complex reception signal vector y of Equation 3, received through the receive antennas 122-1˜122-N, using the reordered left singular vectors (U^(†)=G), as shown in Equation 8:

$\begin{matrix} \begin{matrix} {\overset{\sim}{y} = {{U^{\dagger}{HFs}} + {U^{\dagger}w}}} \\ {{= {{\overset{\_}{\Sigma}\; P_{BL}s} + \overset{\sim}{w}}},} \end{matrix} & \left\lbrack {{Eqn}.\mspace{14mu} 8} \right\rbrack \end{matrix}$

where an N×N diagonal matrix

$\left( {\Sigma \overset{\Delta}{=}{\Lambda \; \Phi}} \right)\mspace{14mu} {is}\mspace{14mu} {diag}{\left\{ {\sigma_{1},\sigma_{N},\sigma_{2},\sigma_{N - 1},\ldots \mspace{14mu},\sigma_{N/2},\sigma_{{N/2} + 1}} \right\}.}$

When a total of N effective channel matrixes are blocked (or grouped) two by two, an i^(th) effective channel submatrix Σ_(i) is defined as

${\Sigma_{i}\overset{\Delta}{=}{{{diag}\left\{ {\sigma_{i},\sigma_{N - i + 1}} \right\} \mspace{14mu} {for}\mspace{14mu} i} = 1}},2,\ldots \mspace{14mu},{N/2},$

and Σ is expressed as Σ=diag{Σ₁, Σ₂, . . . ,Σ_(N/2)}. Therefore, the P_(BL) is expressed as Equation 9:

$\begin{matrix} {P_{BL}\overset{\Delta}{=}{{diag}{\left\{ {P_{1},P_{2},\ldots \mspace{14mu},P_{\frac{N}{2}}} \right\}.}}} & \left\lbrack {{Eqn}.\mspace{14mu} 9} \right\rbrack \end{matrix}$

FIG. 2 is a diagram illustrating a structure of a soft demapper according to an embodiment of the present invention.

Referring to FIG. 2, a soft demapper 200 includes N/2 symbol-by-symbol demappers 200-1˜200-N/2 for symbol-by-symbol detection on real signals, and N/2 symbol-by-symbol demappers 210-1˜210-N/2 for symbol-by-symbol detection on imaginary signals.

A block diagonal matrix ΣP_(BL) obtained by applying the P_(BL) defined in Equation 9 to Equation 8 is as shown in Equation 10:

$\begin{matrix} {{{\begin{bmatrix} {\overset{\sim}{y}}_{1} \\ {\overset{\sim}{y}}_{2} \\ \vdots \\ {\overset{\sim}{y}}_{\frac{N}{2}} \end{bmatrix} = {{\begin{bmatrix} B_{1} & 0 & \ldots & 0 \\ 0 & B_{2} & ⋰ & \vdots \\ \vdots & ⋰ & ⋰ & 0 \\ 0 & \ldots & 0 & B_{\frac{N}{2}} \end{bmatrix}\begin{bmatrix} s_{1} \\ s_{2} \\ \vdots \\ s_{\frac{N}{2}} \end{bmatrix}} + \begin{bmatrix} {\overset{\sim}{w}}_{1} \\ {\overset{\sim}{w}}_{2} \\ \vdots \\ {\overset{\sim}{w}}_{\frac{N}{2}} \end{bmatrix}}},{where}}{{{\overset{\sim}{y}}_{i}\overset{\Delta}{=}\begin{bmatrix} {\overset{\sim}{y}}_{{2\; i} - 1} & {\overset{\sim}{y}}_{2\; i} \end{bmatrix}^{T}},{s_{i}\overset{\Delta}{=}\begin{bmatrix} s_{{2\; i} - 1} & s_{2\; i} \end{bmatrix}^{T}},{and}}{{\overset{\sim}{w}}_{i}\overset{\Delta}{=}{\begin{bmatrix} {\overset{\sim}{w}}_{{2\; i} - 1} & {\overset{\sim}{w}}_{2\; i} \end{bmatrix}^{T}.}}} & \left\lbrack {{Eqn}.\mspace{14mu} 10} \right\rbrack \end{matrix}$

Further, B_(i) denotes a 2×2 subblock matrix of an effective channel and is expressed as a matrix (B_(i) ΔΣ_(i)P_(i)) having only the pure real values.

As a result, the {tilde over (y)} corresponding to an i^(th) subblock is expressed as Equation 11 below for the complex matrix. That is, real parts and imaginary parts of the {tilde over (y)} are decomposed and input to their associated symbol-by-symbol demappers.

$\begin{matrix} {{\begin{bmatrix} {\overset{\sim}{y}}_{i,I} \\ {\overset{\sim}{y}}_{i,Q} \end{bmatrix} = {{\begin{bmatrix} B_{i} & 0 \\ 0 & B_{i} \end{bmatrix}\begin{bmatrix} s_{i,I} \\ s_{i,Q} \end{bmatrix}} + \begin{bmatrix} {\overset{\sim}{w}}_{i,I} \\ {\overset{\sim}{w}}_{i,Q} \end{bmatrix}}},} & \left\lbrack {{Eqn}.\mspace{14mu} 11} \right\rbrack \end{matrix}$

where the subscripts I and Q denote indicators indicating a real part and an imaginary part of a complex matrix, respectively.

Thereafter, the symbol-by-symbol demappers 200-1˜200-N/2 receive 2 consecutive real signals {tilde over (y)}_(i,I), and output vector signals ŝ_(i,I) for the consecutive symbols using the ML equation, or Equation 12 below.

In addition, the N/2 symbol-by-symbol demappers 210-1˜210-N/2 receive 2 consecutive imaginary signals {tilde over (y)}_(i,Q), and output vector signals ŝ_(i,Q) for the consecutive symbols using the ML equation, or Equation 13 below. The symbol-by-symbol demappers 200-1˜200-N2 and 210-1˜210-N/2 independently detect s_(i,I) and s_(i,Q), respectively.

$\begin{matrix} {{\hat{s}}_{i,I} = {\arg \; {\min\limits_{s_{i,I} \in \chi_{\sqrt{M}}^{2}}{{{{\overset{\sim}{y}}_{i,I} - {B_{i}s_{i,I}}}}^{2}.}}}} & \left\lbrack {{Eqn}.\mspace{14mu} 12} \right\rbrack \\ {{\hat{s}}_{i,Q} = {\arg \; {\min\limits_{s_{i,Q} \in \chi_{\sqrt{M}}^{2}}{{{{\overset{\sim}{y}}_{i,Q} - {B_{i}s_{i,Q}}}}^{2}.}}}} & \left\lbrack {{Eqn}.\mspace{14mu} 13} \right\rbrack \end{matrix}$

In conclusion, in the soft demapper 200, since candidate groups of the transmission signals ŝ_(i)=ŝ_(i,I)+j{tilde over (s)}_(i,Q) for i=1,2, . . . ,N/2 acquired through the ML equations of Equation 12 and Equation 13 are equal in number to the constellations of complex symbols used in the transmitter 100, symbol-by-symbol detection is possible.

FIG. 3 is a diagram illustrating an example of an error rate versus SNR of BL-UCD according to an embodiment of the present invention. For example, shown are the simulation results for N_(t)=N_(r)=4 and data rate=4 bps/Hz.

Referring to FIG. 3, it is shown that compared with SVD, BL-UCD is outstandingly lower in a Frame Error Rate (FER) versus SNR.

FIG. 4 is a diagram illustrating another example of an error rate versus SNR of BL-UCD according to an embodiment of the present invention. For example, shown are the simulation results for N_(t)=N_(r)=4 and data rate=6 bps/Hz.

Referring to FIG. 4, it is shown that compared with SVD, BL-UCD is noticeably lower in FER versus SNR.

As is apparent from the foregoing description, the present invention acquires a blockwise channel structure by applying a precoder to the transmitter and a receive filter to the receiver. Thus, during symbol-by-symbol detection on received symbols, the invention acquires the optimal joint-ML performance without error transfer. Therefore, when channel coding, which is widely used in the actual wireless channel environment, is applied, the proposed technology shows improved error probability performance compared with the existing closed-loop technologies. In addition, since the highest-SNR channel and the lowest-SNR channel are grouped into one subblock, it is possible to reduce an SNR difference between all subchannels, approximately to that of existing UCD technology.

Although the present disclosure has been described with an exemplary embodiment, various changes and modifications may be suggested to one skilled in the art. It is intended that the present disclosure encompass such changes and modifications as fall within the scope of the appended claims. 

1. An apparatus for decomposing a channel in a closed-loop Multiple Input Multiple Output (MIMO) communication system, the apparatus comprising: a transmitter for precoding input symbols using a first matrix which is a product of a unitary matrix V, a diagonal matrix Φ and a blockwise Uniform Channel Decomposition (UCD) matrix P_(BL), before outputting the input symbols.
 2. The apparatus of claim 1, wherein the diagonal matrix Φ is a unit matrix.
 3. The apparatus of claim 1, wherein the diagonal matrix Φ is an N×N diagonal matrix; wherein elements φ_(k) of the diagonal matrix Φ are calculated using a water-filling scheme and are expressed as: ${\Phi \overset{\Delta}{=}{{diag}\left\{ {\varphi_{1},\varphi_{N},\varphi_{2},\varphi_{N - 1},\ldots \mspace{14mu},\varphi_{N/2},\varphi_{{N/2} + 1}} \right\}}},{\varphi_{k} = {{\left( {\mu - \frac{\alpha}{\lambda_{k}^{2}}} \right)_{+}^{\frac{1}{2}}\mspace{14mu} {for}\mspace{14mu} k} = 1}},2,\ldots \mspace{14mu},N,$ where μ is a parameter for satisfying a condition of Σ_(k=1) ^(N)φ_(k) ²=N, α denotes an energy ratio of a transmission symbol to noise, ${(a)_{+}\overset{\Delta}{=}{\max \left\{ {0,a} \right\}}},$ λ_(k) is a diagonal element of a diagonal matrix Λ, and Λ=diag{λ₁,λ_(N),λ₂,λ_(N-1), . . . ,λ_(N/2),λ_(N/2+1)}.
 4. The apparatus of claim 3, wherein an element P_(i) of the matrix P_(BL) is expressed as: ${P_{BL}\overset{\Delta}{=}{{diag}\left\{ {P_{1},P_{2},\ldots \mspace{14mu},P_{\frac{N}{2}}} \right\}}},{where}$ ${P_{i} = {{\begin{bmatrix} C_{i} & S_{i} \\ {- S_{i}} & C_{i} \end{bmatrix}\mspace{14mu} {for}\mspace{14mu} i} = 1}},2,\ldots \mspace{14mu},{N/2},{{in}\mspace{14mu} {which}}$ $C_{i} = {\sqrt{\frac{\sqrt{\left( {\sigma_{i}^{2} + \alpha} \right)\left( {\sigma_{N - i + 1}^{2} + \alpha} \right)} - \left( {\sigma_{N - i + 1}^{2} + \alpha} \right)}{\sigma_{i}^{2} - \sigma_{N - i + 1}^{2}}}\mspace{14mu} {and}}$ ${S_{i} = \sqrt{1 - C_{i}^{2}}},{{{where}\mspace{14mu} \sigma_{i}} = {\lambda_{i}{\varphi_{i}.}}}$
 5. The apparatus of claim 1, further comprising: a receiver for generating a reception signal vector by filtering an output signal of the transmitter using a second matrix.
 6. The apparatus of claim 5, wherein the second matrix is a matrix U^(†) obtained by performing conjugate transpose on a singular vector unitary matrix U including a channel matrix H, and the H is expressed as: ${H = {{U\; \Lambda \; V^{\dagger}}\overset{\Delta}{=}{\left\lbrack {u_{1}u_{2}\mspace{14mu} \ldots \mspace{14mu} u_{K}} \right\rbrack {\Lambda \left\lbrack {v_{1}v_{2}\mspace{14mu} \ldots \mspace{14mu} v_{K}} \right\rbrack}^{\dagger}}}},$ where H denotes a channel matrix acquired using a Singular Value Decomposition (SVD) technique, and V denotes a singular value matrix.
 7. The apparatus of claim 6, wherein the reception signal vector {tilde over (y)} is expressed as: ${{\begin{matrix} {\overset{\sim}{y} = {{U^{\dagger}{HF}_{s}} + {U^{\dagger}w}}} \\ {= {{\Sigma \; P_{BL}s} + \overset{\sim}{w}}} \end{matrix}\begin{bmatrix} {\overset{\sim}{y}}_{1} \\ {\overset{\sim}{y}}_{2} \\ \vdots \\ {\overset{\sim}{y}}_{\frac{N}{2}} \end{bmatrix}} = {{\begin{bmatrix} B_{1} & 0 & \ldots & 0 \\ 0 & B_{2} & ⋰ & \vdots \\ \vdots & ⋰ & ⋰ & 0 \\ 0 & \ldots & 0 & B_{\frac{N}{2}} \end{bmatrix}\begin{bmatrix} s_{1} \\ s_{2} \\ \vdots \\ s_{\frac{N}{2}} \end{bmatrix}} + \begin{bmatrix} {\overset{\sim}{w}}_{1} \\ {\overset{\sim}{w}}_{2} \\ \vdots \\ {\overset{\sim}{w}}_{\frac{N}{2}} \end{bmatrix}}},{where}$ ${{\overset{\sim}{y}}_{i}\overset{\Delta}{=}\begin{bmatrix} {\overset{\sim}{y}}_{{2\; i} - 1} & {\overset{\sim}{y}}_{2\; i} \end{bmatrix}^{T}},{s_{i}\overset{\Delta}{=}\begin{bmatrix} s_{{2\; i} - 1} & s_{2\; i} \end{bmatrix}^{T}},{{\overset{\sim}{w}}_{i}\overset{\Delta}{=}\begin{bmatrix} {\overset{\sim}{w}}_{{2\; i} - 1} & {\overset{\sim}{w}}_{2\; i} \end{bmatrix}^{T}},$ and B_(i) denotes a 2×2 subblock matrix of an effective channel and is expressed as a matrix $\left( {B_{i}\overset{\Delta}{=}{\Sigma_{i}\; P_{i}}} \right)$ having only pure real values.
 8. A method for decomposing a channel in a closed-loop Multiple Input Multiple Output (MIMO) communication system, the method comprising: preceding input symbols using a first matrix which is a product of a unitary matrix V, a diagonal matrix Φ and a blockwise Uniform Channel Decomposition (UCD) matrix P_(BL), before outputting the input symbols.
 9. The method of claim 8, wherein the diagonal matrix Φ is a unit matrix.
 10. The method of claim 8, wherein the diagonal matrix Φ is an N×N diagonal matrix; wherein elements φ_(k) of the diagonal matrix Φ are calculated using a water-filling scheme and are expressed as: ${\Phi \overset{\Delta}{=}{{diag}\left\{ {\varphi_{1},\varphi_{N},\varphi_{2},\varphi_{N - 1},\ldots \mspace{14mu},\varphi_{N/2},\varphi_{{N/2} + 1}} \right\}}},{\varphi_{k} = {{\left( {\mu - \frac{\alpha}{\lambda_{k}^{2}}} \right)_{+}^{\frac{1}{2}}\mspace{14mu} {for}\mspace{14mu} k} = 1}},2,\ldots \mspace{14mu},N,$ where μ is a parameter for satisfying a condition of Σ_(k=1) ^(N)φ_(k) ²=N, α denotes an energy ratio of a transmission symbol to noise, (α)₊ Δmax{0, α}, λ_(k) is a diagonal element of a diagonal matrix Λ, and Λ=diag{λ₁,λ_(N),λ₂,λ_(N-1), . . . ,λ_(N/2),λ_(N/2+1)}.
 11. The method of claim 10, wherein an element P_(i) of the matrix P_(BL) is expressed as: ${P_{BL}\overset{\Delta}{=}{{diag}\left\{ {P_{1},P_{2},\ldots \mspace{14mu},P_{\frac{N}{2}}} \right\}}},{where}$ ${P_{i} = {{\begin{bmatrix} C_{i} & S_{i} \\ {- S_{i}} & C_{i} \end{bmatrix}\mspace{14mu} {for}\mspace{14mu} i} = 1}},2,\ldots \mspace{14mu},{N/2},{{in}\mspace{14mu} {which}}$ $C_{i} = {\sqrt{\frac{\sqrt{\left( {\sigma_{i}^{2} + \alpha} \right)\left( {\sigma_{N - i + 1}^{2} + \alpha} \right)} - \left( {\sigma_{N - i + 1}^{2} + \alpha} \right)}{\sigma_{i}^{2} - \sigma_{N - i + 1}^{2}}}\mspace{14mu} {and}}$ ${S_{i} = \sqrt{1 - C_{i}^{2}}},{{{where}\mspace{14mu} \sigma_{i}} = {\lambda_{i}{\varphi_{i}.}}}$
 12. The method of claim 8, further comprising: generating a reception signal vector by filtering the precoded input symbols.
 13. The method of claim 12, wherein the second matrix is a matrix U^(†) obtained by performing conjugate transpose on a singular vector unitary matrix U including a channel matrix H, and the H is expressed as: ${H = {{U\; \Lambda \; V^{\dagger}}\overset{\Delta}{=}{\left\lbrack {u_{1}u_{2}\mspace{14mu} \ldots \mspace{14mu} u_{K}} \right\rbrack {\Lambda \left\lbrack {v_{1}v_{2}\mspace{14mu} \ldots \mspace{14mu} v_{K}} \right\rbrack}^{\dagger}}}},$ where H denotes a channel matrix acquired using a Singular Value Decomposition (SVD) technique, and V denotes a singular value matrix.
 14. The method of claim 13, wherein the reception signal vector {tilde over (y)} is expressed as: ${{\begin{matrix} {\overset{\sim}{y} = {{U^{\dagger}{HF}_{s}} + {U^{\dagger}w}}} \\ {= {{\Sigma \; P_{BL}s} + \overset{\sim}{w}}} \end{matrix}\begin{bmatrix} {\overset{\sim}{y}}_{1} \\ {\overset{\sim}{y}}_{2} \\ \vdots \\ {\overset{\sim}{y}}_{\frac{N}{2}} \end{bmatrix}} = {{\begin{bmatrix} B_{1} & 0 & \ldots & 0 \\ 0 & B_{2} & ⋰ & \vdots \\ \vdots & ⋰ & ⋰ & 0 \\ 0 & \ldots & 0 & B_{\frac{N}{2}} \end{bmatrix}\begin{bmatrix} s_{1} \\ s_{2} \\ \vdots \\ s_{\frac{N}{2}} \end{bmatrix}} + \begin{bmatrix} {\overset{\sim}{w}}_{1} \\ {\overset{\sim}{w}}_{2} \\ \vdots \\ {\overset{\sim}{w}}_{\frac{N}{2}} \end{bmatrix}}},{where}$ ${{\overset{\sim}{y}}_{i}\overset{\Delta}{=}\begin{bmatrix} {\overset{\sim}{y}}_{{2\; i} - 1} & {\overset{\sim}{y}}_{2\; i} \end{bmatrix}^{T}},{s_{i}\overset{\Delta}{=}\begin{bmatrix} s_{{2\; i} - 1} & s_{2\; i} \end{bmatrix}^{T}},{{\overset{\sim}{w}}_{i}\overset{\Delta}{=}\begin{bmatrix} {\overset{\sim}{w}}_{{2\; i} - 1} & {\overset{\sim}{w}}_{2\; i} \end{bmatrix}^{T}},$ and B_(i) denotes a 2×2 subblock matrix of an effective channel and is expressed as a matrix $\left( {B_{i}\overset{\Delta}{=}{\Sigma_{i}\; P_{i}}} \right)$ having only pure real values. 